Current-type GTO inverter with surge energy restoration

ABSTRACT

In a current type GTO inverter, commutation surge voltage is inevitably generated from an inductive load whenever each GTO is turned off. The commutation surge voltage thus generated is once stored in a capacitor (C 1 ) through a diode surge voltage rectifier (5) and then restored to the DC source terminals (3A, 3B) of the GTO bridge-connected inverter (3) through a pair of other GTOs (G 7 , G 8 ) turned on during steady state intervals of inverter commutation. Magnetic energy stored in a reactor (Lr 1 , Lr 2 ) in motor-driving operation is recharged to the capacitor (C 1 ) through the diode surge voltage rectifier (5) after the GTOs (G 7 , G 8 ) have been turned off; the motor kinetic energy stored in the capacitor (C 1 ) through diodes (D 8 , D 9 ) in motor-braking operation is regenerated to the AC source side of the inverter (3) through a pair of other GTOs (G 9 , G 10 ) when the voltage across the capacitor (C 1 ) exceeds a predetermined value, and magnetic energy stored in the reactor (Lr 1 , Lr 2 ) in motor-braking operation is recharged to the capacitor (C 1 ) through diodes (D 12 , D 13 ) after the GTOs (G 9 , G 10 ) have been turned off. The circuit operation is stable at higher frequency range because no vibration circuits are provided, and the energy conversion efficiency is high because every energy loss is effectively restored to the inverter or the power source side.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates generally to a current type GTO (gate turn-off thyristor) inverter, and more specifically to a surge voltage clamping circuit for clamping the surge voltage generated when each GTO incorporated in a current type GTO bridge-connected inverter is turned off. The clamped surge voltage is stored once in a capacitor and then returned to the terminals between the rectifier and the GTO inverter for energy restoration.

2. Description of the Prior Art

In a current-type gate turn-off thyristor bridge-connected inverter, since gate turn-off thyristors (referred to as GTOs, simply hereinafter) are incorporated in the inverter as the main switching elements, no commutating circuit including a commutation reactor, for instance, is required, because the GTO can be turned from on to off or vice versa in response to a control signal applied to the gate terminal thereof. Here, the terminology "commutation" means that the load current of one phase is switched to that of another plane or vice versa by thyristor switching operation. In the above-mentioned current-type GTO inverter, however, in the case where a load such as an induction motor having an inductance is coupled, commutation surge voltages are inevitably generated whenever each GTO is turned off. The generated surge voltages are superimposed upon the alternating output voltage of the GTO inverter, thus resulting in a problem in that some of the GTOs may be damaged by these commutation surge voltages.

In order to overcome the above problem, a commutation surge voltage clamping circuit has been proposed, by which the commutation surge voltages generated whenever each GTO is turned off are absorbed or stored in a single electrolytic capacitor and thereafter returned to the load side through the GTO inverter for reducing the electric power loss. This function is called energy restoration.

In the conventional commutation surge voltage clamping circuit used for a current type GTO inverter, however, there exist some disadvantages as follows.

(1) Since a pair of ordinary thyristors are used for restoring the stored commutation surge voltage energy to the DC source terminals of the GTO inverter, two vibration circuits or thyristor turning-off circuits including a capacitor and an inductor respectively are necessary. Further, since the surge voltage energy is restored through these capacitors used for the vibration circuits, the capacity of these capacitors of the vibration circuits is determined to be relatively large. As a result, the turn-off operation of the ordinary thyristors often fails at higher frequency range. In other words, it is impossible to stably operate the commutation surge voltage clamping circuit when the GTO inverter operates at a high speed.

(2) Since the commutation surge voltage energy is restored from the electrolytic capacitor to the DC source terminals of the GTO inverter through the vibration capacitors connected in series with the electrolytic capacitor, the capacitance of the restoring circuit is relatively large. Therefore, a reactor having a large inductance is necessary in order to smooth the current restored to the GTO inverter. In other words, the cost of the commutation surge voltage clamping circuit is relatively high.

(3) Since the charging and discharging circuits of the capacitor are operable only when the motor is driven in the forward or the reverse direction, when the motor is being braked, it is impossible to regenerate the motor kinetic energy stored in the capacitor in motor-braking operation to the AC source side of the inverter or to charge the magnetic energy stored in the reactor in motor-braking operation in the capacitor.

A more detailed description of the prior-art commutation surge voltage clamping circuit will be made with reference to the attached drawings under DESCRIPTION OF THE PREFERRED EMBODIMENT.

SUMMARY OF THE INVENTION

With these problems in mind, therefore, it is a primary object of the present invention to provide a surge voltage clamping circuit for a current type GTO inverter which can operate stably at high frequency range.

It is another object of the present invention to provide a surge voltage clamping circuit for a current type GTO inverter in which no vibration circuit for turning off the energy-restoring thyristor is provided without use of a large-inductance reactor, and therefore the circuit configuration is simplified or reducing the manufacturing cost.

It is still the other object of the present invention to provide a surge voltage clamping circuit for a current type GTO inverter which can also regenerate the motor kinetic energy while the motor is driven in the forward or reverse direction or being braked.

To achieve the above-mentioned object, a surge voltage clamping circuit for a current type GTO inverter according to the present invention comprises (a) a GTO bridge-connected inverter, (b) a thyristor bridge-connected rectifier, (c) a diode bridge-connected commutation surge voltage rectifier, (d) a capacitor for storing commutation surge voltage energy, (e) a cumulative reactor, (f) a DC reactor, (g) a first GTO, (h) a second GTO, (i) a first diode, (j) a second diode, (k) a third GTO, (l) a fourth GTO, (m) a third diode and (n) a fourth diode. In the circuit configuration thus constructed, commutation surge voltage energy stored in said capacitor in motor-driving operation is restored to said inverter through said first and second GTOs during steady state intervals of inverter commutation; magnetic energy stored in said reactor in motor-driving operation is recharged to said capacitor through said first and second diodes after said first and second GTOs have been turned off; the motor kinetic energy stored in said capacitor in motor-braking operation is regenerated to the AC source through said third and fourth GTOs when the voltage across said capacitor exceeds a predetermined value; and magnetic energy stored in said reactor in motor-braking operation is recharged to said capacitor through said third and fourth diodes after said third and fourth GTOs have been turned off.

BRIEF DESCRIPTION OF THE DRAWINGS

The features and advantages of a surge voltage clamping circuit for a current-type GTO inverter according to the present invention over the prior art clamping circuit will be more clearly appreciated from the following description of the preferred embodiment of the invention taken in conjunction with the accompanying drawings in which like reference numerals designate the same or similar elements or sections throughout the figures thereof and in which:

FIG. 1 is a circuit diagram of a prior-art surge voltage clamping circuit for a current-type GTO inverter;

FIG. 2 is a timing chart of the prior-art surge voltage clamping circuit for the current-type GTO inverter shown in FIG. 1, for assistance in explaining the operation thereof;

FIG. 3A is a coordinate with rotor angular frequency as abscissa and with rotor torque as ordinate, for assistance in explaining four quadrant operation of one feature of the present invention;

FIG. 3B is a table showing signs of the rotor torque and the rotor angular frequency being classified into four quadrants;

FIG. 3C is a graphical representation showing an example of the four-quadrant operation, in which motor driving mode is shifted from the first quadrant operation, through the second quadrant operation, to the third quadrant operation when the reference rotor angular frequency is switched from the normal direction to the reverse direction; and

FIG. 4 is a circuit diagram of the surge voltage clamping circuit for a current-type GTO inverter according to the present invention;

FIG. 5(A) is a partial circuit diagram of the surge voltage clamping circuit for the current-type GTO inverter according to the present invention, in which a steady-state current path obtained when GTO G₁ and GTO G₆ are both turned on is depicted by thicker lines;

FIG. 5(B) is the same partial circuit diagram as in FIG. 5(A), in which transient-state current paths obtained immediately after GTO G₁ has been turned off and GTO G₃ has been turned on with GTO G₆ kept on are depicted by thicker lines;

FIG. 6 is a partial circuit diagram of the surge voltage clamping circuit according to the present invention, in which only the charging and discharging paths of the capacitor C₁ are shown;

FIG. 7(A) is a partial circuit diagram of the surge voltage clamping circuit according to the present invention, through which surge voltage energy stored in the capacitor is discharged or restored to the inverter in motor-driving operation;

FIG. 7(B) is a similar partial circuit diagram, through which magnetic energy stored in the reactor is charged into the capacitor in motor-driving operation;

FIG. 7(C) is a similar partial circuit diagram, through which motor kinetic energy stored in the capacitor is discharged or regenerated to the AC source side in motor-braking operation;

FIG. 7(D) is a similar partial circuit diagram, through which magnetic enegy stored in the reactor is charged into the capacitor in motor-braking operation;

FIG. 8 is a graphical representation showing the relationship between the off-time interval τ of GTOs G₇ and G₈ and the capacitor voltage e_(c1) with the capacitor C₁ as parameter at a fixed dc source voltage Ed and a fixed inverter frequency f;

FIG. 9 is a graphical representation showing the relationship between the inverter frequency f and the capacitor voltage e_(c1) with the off-time interval τ as parameter at a fixed dc source voltage Ed and a fixed capacitor C₁ ;

FIG. 10 is a graphical representation showing the relationship between the dc source voltage Ed and the capacitor voltage e_(c1) at a fixed inverter frequency f, a fixed off-time interval τ and a fixed capacitor C₁ ;

FIG. 11(A) is a equivalent circuit diagram corresponding to the current path shown by thicker lines in FIG. 5(A), in which GTOs G₁ and G₆ are both on and the circuit is in the steady state;

FIG. 11(B) is a equivalent circuit diagram corresponding to the current paths shown by thicker lines in FIG. 5(B), in which GTO G₁ has been turned off and GTO G₃ has been turned of with GTO G₆ kept on and the circuit is in the transient state; and

FIG. 12 is a timing chart of the surge voltage clamping circuit for the current-type GTO inverter shown in FIG. 4, in which the time interval between two dashed lines partially corresponds to that shown in FIG. 2.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT

To facilitate understanding of the present invention, a reference will be made to an example of prior-art surge voltage clamping circuits for a current type GTO bridge-connected inverter, with reference to the attached drawings.

With reference to FIG. 1, the GTO inverter provided with a surge voltage clamping circuit is roughly made up of a thyristor bridge-connected current rectifier 1, a DC reactor having two inductive reactances 2A and 2B magnetically connected each other, a GTO (gate turn-off thyristor) bridge-connected inverter 3, an induction motor 4, a diode bridge-connected commutation surge voltage rectifier 5, and a commutation surge voltage clamping circuit 6 including an electrolytic capacitor C₁. The above clamping circuit 6 functions also as a commutation surge voltage energy restoring circuit.

The thyristor bridge-connected rectifier 1 includes six rectifying thyristors RT₁ to RT₆, which are turned on sequentially in the order of RT₁ and RT₆, RT₃ and RT₂ and RT₅ and RT₄ so that each half cycle of the sine wave of a three-phase power source can be passed in sequence. Therefore, when the rectifying thyristors RT₁ and RT₆ are both turned on, a first-phase current R is supplied from the terminal A to the terminal C by way of thyristor RT₁, reactor 2A, terminal 3A, GTO bridge-connected inverter 3, motor 4, GTO bridge-connected inverter 3, terminal 3B, reactor 2B, and thyristor RT₆. Similarly, when the rectifying thyristors RT₃ and RT₂ are both turned on, a second-phase current S is supplied from the terminal B to the terminal A by way of rectifying thyristor RT₃, reactor 2A, terminal 3A, GTO bridge-connected inverter 3, motor 4, GTO bridge-connected inverter 3, terminal 3B, reactor 2B and the rectifying thyristor RT₂ ; when the rectifying thyristors RT₅ and RT₄ are turned on, a third phase current T is supplied from the terminal C to the terminal B by way of the rectifying thyristor RT₅, reactor 2A, terminal 3A, GTO bridge-connected inverter 3, motor 4, GTO bridge-connected inverter 3, terminal 3B, reactor 2B, and rectifying thyristor RT₄. The rectified full-wave direct current is further smoothed through the DC cumulative reactor having two inductances 2A and 2B magnetically connected each other. Therefore, the smoothed direct current Id is further converted into an alternate current of an appropriate frequency through the GTO inverter 3 to drive the induction motor 4 at any desired speed.

The GTO inverter 3 includes six bridge-connected GTOs G₁ to G₆. When the GTOs are turned on in the order of G₁ and G₆ and then G₃ with G₆ on in sequence for each 60 degrees, an alternate square-wave U-phase current i_(u) and a V-phase current i_(v) with each pulse width of 60 degrees is first obtained by the GTO inverter. When the GTOs are turned on in the order of G₃ and G₆ and then G₂ with G₃ on in sequence for each 60 degrees, an alternate square-wave V-phase current i_(v) with each pulse width of 120 degrees is obtained by the GTO inverter. When the GTOs are turned on in the order of G₃ and G₂ and then G₅ with G₂ on in sequence for each 60 degrees, an alternate square-wave V-phase current i_(v) and a W-phase current i_(w) with each pulse width of 60 degrees is obtained by the GTO inverter.

In other words, when GTOs G₁ and G₆ are turned on, the rectified direct current Id flows as the U-phase current (the latter half of 120 degrees) i_(u) through the first U-phase winding having reactance X_(u) and the third W-phase winding having reactance X_(w) of the motor. When GTO G₁ is turned off and GTO G₃ is turned on with GTO G₆ kept turned on, the current Id flows as the V-phase current (the first half of 120 degrees) i_(v) through the second V-phase winding having reactance X_(v) and the third W-phase winding having reactance X_(w).

Similarly, when G₆ is turned off and G₂ is turned on with G₃ kept turned on, the current Id flows as the V-phase current (the latter half of 120 degrees) i_(v) through X_(v) and X_(u). When G₃ is turned off and G₅ is turned on with G₂ kept turned on, the current Id flows as the W-phase current (the first half of 120 degrees) i_(w) through X_(w) and X_(u).

In summary, GTOs are turned on or off in the order of G₁, G₆, G₃, G₂, G₅ and G₄ for each 60 degrees. The current passed through these three-phase windings having motor reactances X_(u), X_(v), and X_(w) generates a rotational magnetic flux.

Whenever each of these three-phase windings is switched off, commutation surge voltages are inevitably generated by the magnetic energy stored in the respective windings. In order to absorb these commutation surge voltages, there is additionally provided a surge voltage clamping circuit including a diode bridge-connected commutation surge voltage rectifying section 5 made up of fix diodes D₁ to D₆ and a surge voltage clamping section 6 made up of an electrolytic capacitor C₁, two capacitors C₂ and C₃, two ordinary thyristors T₁ and T₂, four diodes D₈ to D₁₁, two turning-off inductors L₁ and L₂, and a reactor having two reactances Lr₁ and Lr₂, as shown in FIG. 1. In the above reactor, the positive side of the first reactance Lr₁ is connected to the positive terminal of the DC rectifier 1 through the diode D₁₀ ; the negative side of the second reactance Lr₂ is connected to the negative terminal of the DC rectifier through the diode D₁₁, respectively.

The operation of the prior-art surge voltage clamping circuit will be described hereinbelow with reference to FIGS. 1 and 2.

When mode I (G₁ and G₆ are on) is switched to mode II (G₁ is off, G₆ is on, G₃ is on), for instance, as depicted in FIG. 2, the current i_(u) flowing through windings X_(u) and X_(w) is commutated to the current i_(v) flowing through windings X_(v) and X_(w). In this transient state, the current i_(u) does not immediately fall to zero level but decreases gradually and the current i_(v) does not immediately rise to the current Id but increases gradually as depicted in FIG. 2. This is because there exists each inductance in each winding and thereby an induced surge voltage is inevitably generated across each winding. It is very important to suppress or eliminate these induced surge voltages for protection of GTO thyristors.

An induced surge voltage V_(vw) developed across the windings X_(v) and X_(w) in this transient state can be charged in the capacitor C₁ as follows: When the surge voltage V_(vw) exceeds the voltage across the capacitor C₁, since the diodes D₃ and D₆ are both forward biased (the anode of D₃ is high in voltage level; the cathode of D₆ is low in voltage level), the major part of the current to be passed through the winding X_(v) flows by way of GTO G₃, diode D₃, capacitor C₁, diode D₆ and GTO G₆. In this transient state, the surge voltage V_(vw) is suppressed by the capacitor C₁ if the voltage e_(c1) across the capacitor C₁ is sufficiently low.

Simultaneously, when an induced surge voltage V_(uv) developed across the windings X_(u) and X_(v) in this transient state falls to the voltage e_(c1) of the capacitor C₁, since the diode D₃ and D₂ are both forward biased (the cathode of D₂ is low in voltage level and anode of D₃ is high in voltage level), the commutation energy generated across the windings X_(u) and X_(v) is charged into the capacitor C₁ by way of GTO G₃, diode D₃, capacitor C₁, diode D₂, winding X_(u), winding X_(w) and GTO G₆. In this transient state, the surge voltage V_(uv) is suppressed by the capacitor C₁. As a result, the induced surge voltage V_(wu) developed across the windings X_(w) and X_(u) becomes zero as shown in FIG. 2. The current i_(v) increases gradually up to the direct current Id in accordance with a time constant determined by the circuit constant of the motor load. When the current i_(u) reaches zero, the diode D₂ is cut off. Simultaneously, no induced surge voltage is generated in the winding X_(u). When the induced surge voltage V_(vw) falls below the capacitor voltage e_(c1), the diodes D₃ and D₆ are both cut off, so that the capacitor C₁ is electrically disconnected from the inverter 3 and thus the commutation from GTO G₁ to GTO G₃ is completed.

The above-mentioned mode II corresponds to the overlapped (transient) period in a series-connected diode type current inverter. However, there still exists a difference between the GTO inverter shown in FIG. 1 and the series-connected diode type current inverter in that two transient currents flow through the each-phase winding in the directions opposite to each other being superimposed upon each other.

When the charged-up voltage e_(c1) in the capacitor C₁ increases sufficiently, the two reverse blocking ordinary thyristors T₁ and T₂ are turned on in response to a pulse applied to each gate terminal. Therefore, the surge voltage energy stored in the capacitor C₁ is discharged to the DC source terminals 3A and 3B of the GTO inverter 3 by way of inductor L₁, thyristor T₁, reactor Lr₁, diode D₁₀, reactor 2A, GTO inverter 3, motor 4, GTO inverter 3, reactor 2B, diode D₁₁, reactor Lr₂, thyristor T₂, and inductor L₂. The above-mentioned discharge is called commutation surge voltage energy restoration or energy rebound. In this initial state of discharge, the capacitors C₂ and C₃ are also charged up with the polarity as shown in FIG. 1. These two capacitors C₂ and C₃ form two vibration circuits independently. The first vibration circuit is made up of the capacitor C₂ and the inductor L₁ ; the second vibration circuit is made up of the capacitor C₃ and the inductor L₂, each having a relatively high frequency or a relatively small time constant. An example of the voltage wave form e_(c2) or e_(c3) across the capacitor C₂ or C₃ is also shown in FIG. 2. When the polarity of the capacitor C₂ or C₃ of the vibration circuit is reversed, the thyristor T₁ or T₂ is automatically turned off, because a positive potential is applied to the cathode of the hyristor T₁ to T₂. In this state, it should be noted that the polarity of the charged-up voltages of the three capacitors C₁, C₂, C₃ are the same, that is, three charged-up voltages are added to each other. Therefore, when the addition of these three charged-up voltages exceeds the DC source voltage across the thyristor bridge rectifier 1, the energy stored in these three capacitors C₁, C₂ and C₃ are returned to the DC source terminals 3A and 3B by way of the reactor Lr₁, diode 10, reactor 2A, GTO inverter 3, motor 4, GTO inverter 3, reactor 2B, diode D₁₁ and the reactor Lr₂. In this energy restoration operation, since the inductance Lr₁ or Lr₂ of the reactor is so determined as to be sufficiently greater than that of the turning-off (vibration) inductor L₁ or L₂, the two vibration circuits L₁ ·C₂ and L₂ ·C₃ can stably vibrate and provide an sufficient turn-off time for the thyristor T₁ or T₂. In other words, the reactor Lr₁ or Lr₂ functions as a smoothing element. After the surge voltage energy has been discharged, the capacitors C₂ and C₃ are charged again in the direction as shown in FIG. 1, to the voltage level roughly the same as the voltage e_(c1) across the capacitor C₁, because three capacitors C₁, C₂, and C₃ are connected in series.

In this state, since the capacitance of capacitor C₁ is determined to be sufficiently great as compared with that of the capacitors C₂ or C₃, after the capacitors C₂ and C₃ have been charged up, the magnetic energy stored in the reactor Lr₁ or Lr₂ is recharged into the capacitor C₁ by way of diode D₁₁, reactor Lr₂, diode D₈, capacitor C₁, diode D₉, reactor Lr₁ and diode D₁₀. This energy is unavailable reactive power by nature. After the reactor energy has been recharged into the capacitor C₁, the two diodes D₁₀ and D₁₁ are both returned to its off state, respectively, that is, to the initial conditions. Therefore, the electric discharge of the capacitor C₂ or C₃ is prevented for being ready for the succeeding commutation of the GTO bridge-connected inverter 3. After the GTO G₁ has been turned off and the GTO G₃ has been turned on, that is, the commutation has been completed from G₁ to G₃, the operation mode shifts to the mode III in which the driving current is supplied from GTO G₃, through windings X_(v) and X_(w), to GTO G₆.

In the prior-art surge voltage clamping circuit for the current-type GTO inverter described above, however, there exist some disadvantages as follows.

(1) The turning-off inductors L₁ and L₂ and the turning-off capacitors C₁ and C₂ are required for forming two vibration circuits in order to turn off the ordinary thyristors T₁ and T₂ after the surge voltage energy stored in the capacitor C₁ has been restored to the DC source terminals of the GTO inverter 3. Additionally, the commutation energy is restored to the GTO inverter 3 mainly through the capacitors C₂ and C₃. Therefore, in order to sufficiently restore the stored surge voltage energy even under a heavy load, the capacitance of C₂ or C₃ should be relatively large. When capacitors having a large capacitance are used, the vibration frequency becomes low, thus resulting in turn-off failure of the ordinary thyristors T₁ and T₂. In other words, it is impossible to stably operate the surge voltage clamping circuit at a high frequency range when a heavy load is applied to the induction motor.

(2) Since the added charged-up voltage of the series-connected capacitors C₁, C₂, and C₃ is restored to the GTO inverter 3 through the cumulative reactor having reactances Lr₁ and Lr₂, a large inductance is required for this reactor. Otherwise, current overshoot may be generated. In other words, the cost of the reactor Lr₁ and Lr₂ is relatively high.

By the way, in order to drive an induction motor in the same manner as in a DC motor, four-quadrant operation is indispensable. This four-quadrant operation will be described below. As depicted in FIG. 3A, when the rotor angular frequency ω_(r) is taken as abscissa and the motor torque T is taken as ordinate, the first quadrant indicates that a motor is driven in the normal rotational direction; the second quadrant indicates that the motor is braked while rotating in the normal direction; the third quadrant indicates that the motor is driven in the reverse rotational direction; the fourth quadrant indicates that the motor is braked while rotating in the reverse direction.

In other words, in the first quadrant, the motor torque T is positive and the rotor angular frequency ω_(r) also is positive; in the second quadrant, T is negative but ω_(r) is positive; in the third quadrant, T is negative and ω_(r) is also negative; in the fourth quadrant, T is positive but ω_(r) is negative, as depicted in FIG. 3B.

FIG. 3C shows an example in which a motor rotating in the normal direction is switched to the reverse direction at time t₁. In more detail, when a motor is rotating in the 1st quadrant operation (T>0, ω_(r) >0), if the reference frequency (speed) +w*_(r) is switched to -w*_(r), the motor rotates in the 2nd quadrant operation (T<0, ω_(r) >0) (the motor is braked or the motor torque is adsorbed). The instant the rotor frequency reaches zero, the motor begins to rotate in the 3rd quadrant operation (T<0, ω_(r) <0) (the motor is driven in the reverse direction).

Further, in the above description, it should be noted that while the induction motor is being braked for stopping the motor or for reversing the rotational direction of the motor, the induction motor operates as a generator which can return the motor rotational kinetic energy to the AC source side.

Therefore, the prior-art surge voltage clamping circuit shown in FIG. 1 has the following third disadvantages:

(3) Since the charging and discharging circuits for the capacitor is provided only for motor-driving operation, it is impossible to discharge or regenerate the motor kinetic energy stored in the capacitor in motor-braking operation (motor operates as a generator in brake) to the AC source side or to charge the magnetic energy stored in the reactor in motor-braking operation in the capacitor.

In view of the above description, reference is now made to an embodiment of a surge voltage clamping circuit for a current-type GTO inverter according to the present invention with reference to FIG. 4. In this embodiment, GTOs are incorporated in the surge voltage clamping circuit, without providing turning-off (vibration) circuits, in order to operate the circuit stably at a high speed.

In FIG. 4, the points different from the prior-art surge voltage clamping circuit shown in FIG. 1 are that (1) a single direct-current reactor 7 is incorporated in place of the cumulative direct-current reactors 2A and 2B and (2) four gate turn-off thyristors (GTOs) G₇, G₈, G₉ and G₁₀ and four diodes D₈, D₉, D₁₂ and D₁₃ are incorporated without providing the vibration circuits including two inductors L₁ and L₂ and capacitors C₂ and C₃.

A surge voltage clamping circuit 6 or a surge energy restoring circuit according to the present invention comprises an electrolytic capacitor C₁ for absorbing the commutation surge voltage energy, four GTOs G₇, G₈, G₉ and G₁₀ for restoring the surge voltage energy stored in the capacitor C₁ to the GTO inverter 3, a cumulative reactor having two inductive reactances Lr₁ and Lr₂ magnetically connected to each other for smoothing the current restored from the capacitor C₁ to the GTO inverter 3, and four diodes D₈, D₉, D₁₂, and D₁₃ for transferring the magnetic energy stored in the reactor Lr₁, Lr₂ to the capacitor C₁ after the surge voltage energy stored in the capacitor C₁ has been restored.

The first pair of GTOs G₇ and G₈ serve to discharge or restore the surge voltage energy generated in motor-driving operation to the DC source terminals of the GTO inverter 3; the second pair of GTOs G₉ and G₁₀ serve to discharge or regenerate the motor kinetic energy generated in motor braking operation to the AC source side of the GTO inverter 3.

The electrolytic capacitor C₁ is connected in parallel with the diode bridge-connected commutation surge voltage rectifier 4. The first two GTOs G₇ and G₈ are connected between the reactor Lr₁ and Lr₂ and the capacitor C₁ as follows: the positive side of the first winding Lr₁ is connected to the positive terminal of the DC source; the negative side of the first winding Lr₁ is connected to the cathode of the first GTO G₇ ; the anode of the first GTO G₇ is connected to the positive side of the capacitor C₁ ; the negative side of the second winding Lr₂ is connected to the negative terminal of the DC source; the positive side of the second winding Lr₂ is connected to the anode of the second GTO G₈, and the cathode of the second GTO G₈ is connected to the negative side of the capacitor C₁, respectively, respectively.

Further, the two diodes D₈ and D₉ are connected between the reactor Lr₁ and Lr₂ and the capacitor C₁ as follows: the cathode of the first diode D₈ is connected to the positive side of the capacitor C₁ ; the anode of the first diode D₈ is connected to the positive side of the second winding Lr₂ ; the cathode of the second diode D₉ is connected to the negative side of the first winding Lr₁ and the anode of the second diode D₉ is connected to the negative side of the capacitor C₁.

The additional second two GTOs G₉ and G₁₀ for regenerating the motor rotational energy produced in motor braking operation to the AC source side are connected between the reactor Lr₁ and Lr₂ and the capacitor C₁ as follows: the anode of the third GTO G₉ is connected to the positive side of the capacitor C₁ ; the cathode of the third GTO G₉ is connected to the positive side of the second winding Lr₂, the anode of the fourth GTO G₁₀ is connected to the negative side of the first winding Lr₁ ; and the cathode of the fourth GTO G₁₀ is connected to the negative side of the capacitor C₁.

Further, additional two diodes D₁₂ and D₁₃ are connected between the reactor Lr₁ and Lr₂ and the capacitor C₁ as follows: the anode of the third diode D₁₂ is connected to the negative side of the first winding Lr₁ ; the cathode of the third diode D₁₂ is connected to the positive side of the capacitor C₁ ; the anode of the fourth diode D₁₃ is connected to the negative side of the capacitor C₁ ; and the cathode of the fourth diode D₁₃ is connected to the positive side of the second winding Lr₂.

In order to distinguish between the four GTOs and four diodes, the GTOs G₇ and G₈ are referred to as in-drive energy restoring GTOs; the GTOs G₉ and G₁₀ are referred to as in-brake energy regenerating GTOs; the diodes D₈ and D₉ are referred to as in-drive energy restoring diodes; the diodes D₁₂ and D₁₃ are referred to as in-brake energy regenerating diodes, hereinafter.

As already described with reference to FIG. 2, when the GTO G₁ is turned off and the GTO G₃ is turned on with the GTO G₆ kept turned on, the current i_(v) cannot rise immediately and the current i_(u) cannot fall to zero immediately because a commutation surge voltage is developed. The time interval within which two currents i_(u) and i_(v) overlap each other corresponds to a transient state. The time interval within which a single current i_(u) or i_(v) exists corresponds to a steady state.

The above-mentioned energy restoring GTOs G₇ and G₈ are turned on only in the driving state and GTOs G₉ and G₁₀ are turned on only in the braking state but turned off in the transient state, in response to each gate signal generated by each gate circuit (not shown), in order to realize the energy restoration or regeneration function.

The operation of the embodiment of the surge voltage clamping circuit for a current-type GTO inverter according to the present invention will be described hereinbelow with reference to FIGS. 5(A), 5(B), 6, 7(A) to 7(D). Further, the operation is described only during a one-sixth period (60 degrees) of the inverter and during the commutation from GTO G₁ (U phase) to GTO G₃ (V phase).

FIG. 5(A) shows a steady state (single current period) where GTOs G₁ and G₆ are both turned on, so that a load current Id flows from the positive terminal P to the negative terminal N by way of DC reactor 7, GTO G₁, reactance X_(u), reactance X_(w) and GTO G₆ as a U-phase constant load current i_(u) as depicted in FIG. 2.

In general, since no time delay exists due to inductive elements in the gate circuit of GTOs, it is possible to instantaneously turn off GTO G₁ and to instantaneously turn on GTO G₃ in response to gate signals. However, the instant GTO G₁ is turned off and GTO G₃ is turned on with GTO G₆ kept on, a transient state occurs in the inverter circuit 3 as shown in FIG. 2 and FIG. 5(B), in which both the U-phase current i_(u) and the V-phase current i_(v) flow (overlap current period). In more detail, upon turning-on of GTO G₃, the V-phase current i_(v) to be passed through the reactances X_(v) and X_(w) cannot rise immediately due to the presence of the inductance, as depicted in FIG. 2. When this transient induced surge voltage V_(vw) developed across the induction motor reactances X_(v) and X_(w) (positive at X_(v) and negative at X_(w)) is applied to the capacitor voltage e_(c1), since the diodes D₃ and D₆ are both forward-biased, the major part of current to be passed through the reactance X_(v) is bypassed by way of GTO G₃, diode D₃, capacitor C₁, diode D₆ and GTO G₆. However, this transient induced surge voltage V_(vw) is charged into the capacitor C₁ when the voltage across the capacitor C₁ is sufficiently low.

Simultaneously, another transient induced surge voltage V_(uv) is developed across the induction motor reactances X_(u) and X_(u) (positive at X_(v) and negative at X_(u)). While this surge voltage V_(uv) rises up to the capacitor voltage e_(c1), since the diodes D₃ and D₂ are both forward-biased, the surge voltage V_(uv) is restored to the capacitor C₁ by way of diode D₃, capacitor C₁, diode D₂, reactance X_(u), reactance X_(w), and GTO G₆. As a result, the surge voltage V_(wu) across the motor reactances X_(w) and X_(u) is reduced to zero. The V-phase current i_(v) gradually increases up to the direct current Id in accordance with a time constant determined by the circuit constants dependent upon the capacitor voltage e_(c1) at that moment. When the U-phase current i_(u) reaches zero, the diode D₂ is off; the induced surge voltage V_(vw) is no longer produced lower than the capacitor voltage e_(c1), so that the diodes D₃ and D₆ are both off. In this state, the capacitor C₁ is isolated perfectly from the GTO bridge-connected inverter 3, thus the commutation from GTOs G₁ to G₃ being completed with the GTO G₆ kept turned on.

FIG. 6 shows charging and discharging paths of the capacitor C₁ in the surge voltage clamping circuit 6 shown in FIG. 4, by which it is possible to better understand the operation to charge energy to the capacitor C₁ or the operation to discharge the charged energy to the GTO inverter 3.

Further, these charging and discharging paths shown in FIG. 6 are separately depicted in FIGS. 7(A) to 7(D), being classified into four states.

(1) Surge energy restoration in motor-driving operation:

Since the GTOs G₇, G₈ and G₉, G₁₀ are all turned off in the transient interval (commutation period or overlap current period), the surge voltage energy is charged as pulsive current i_(DBR) (shown in FIG. 12) from the inverter 3 to the capacitor C₁ through the diode bridge-connected rectifier 5 as already described with reference to FIG. 5(B), so that the capacitor voltage e_(c1) increases gradually. When the energy restoring GTOs G₇ and G₈ are turned on in the steady state interval (single current period), the energy stored in the capacitor C₁ is discharged (restored) to the DC source terminals 3A and 3B of the GTO inverter 3 by way of capacitor C₁, GTO G₇, reactor Lr₁, reactor 7, GTO inverter 3, motor 4, GTO inverter 3, reactor Lr₂, GTO G₈, and capacitor C₁, as depicted in FIG. 7(A).

Further, since the two GTOs G₇ and G₈ are kept turned off in the transient state interval, the magnetic energy stored in the reactor Lr₁, Lr₂ is charged into the capacitor C₁ by way of reactor Lr₂, diode D₈, capacitor C₁, diode D₉, reactor Lr₁, as depicted in FIG. 7(B). Therefore, the capacitor C₁ is charged up in the same porality as in the surge voltage.

(2) Motor kinetic energy regeneration in motor-braking operation:

When the induction motor is braked, the frequency of the inverter current is lowered; that is, each phase of gate signals applied to the gate terminals of the GTO inverter 3 is delayed from the motor speed. As a result, the polarity of the voltage generated from the inverter 3 is reversed because the motor operates as a generator. In other words, the direction of the current flowing through the reactor Lr is reversed. However, the polarity of the capacity C₁ should be kept in a predetermined direction.

The voltage generated by the motor kinetic energy is charged into the capacitor C₁ by way of reactor Lr₂, diode D₈, capacitor C₁, diode D₉ and reactor Lr₁ as depicted in FIG. 7(B). In this state, it should be noted that regenerating GTOs G₉, G₁₀ are both reverse biased by the diodes D₈ and D₉. Under these conditions, these GTOs will not be turned on even if a gate signal is applied to each of the gate terminals. However, after the current supplied from the reactor Lr₁ and Lr₂ to the capacitor C₁ has decreased to near zero, the energy regenerating GTOs G₉, G₁₀ can be turned on, so that the energy charged in the capacitor C₁ is discharged or regenerated to the AC source side by way of capacitor C₁, GTO G₉, reactor Lr₂, the thyristor bridge-connected current rectifier 1, reactor LR₁, GTO G₁₀ and capacitor C₁ as depicted in FIG. 7(C). In this state, since the voltage e_(c1) is high, the diodes D₁₂ and D₁₃ are reverse biased. However, since the regeneration energy in motor braking operation becomes great in a moment and therefore the capacitor C₁ is immediately charged up. It is preferable to turn on the GTO G₉ and G₁₀ whenever the voltage e_(c1) across the capacitor C₁ exceeds a predetermined reference value.

Further, when the voltage e_(c1) drops below a predetermined value and therefore the GTOs G₉, G₁₀ are turned off, the magnetic energy stored in the reactor Lr in motor-braking operation is recharged into the capacitor C₁ by way of reactor Lr₁, diode D₁₂, capacitor C₁, diode D₁₃, and reactor Lr₂ as depicted in FIG. 7(D). The above-mentioned regenerative braking operation is completed while the motor is being braked. When the induction motor is switched from braking operation to reverse operation, the phase order of the GTO inverter 3 is reversed, as compared with that when the induction motor is driven in the forward direction. In this case, since only the phase order is reversed, the operation is quite the same as that when the motor is driven in the forward direction as shown in FIGS. 7(A) and 7(B).

As described above, the surge voltage clamping circuit of the present invention provides the first function to absorb or charge the surge voltage energy generated in transient state interval (commutation period) and to restore or discharge the surge voltage energy to the DC source terminal of the GTO inverter in the steady state interval in the motor-driving operation, the second function to absorb or charge the magnetic energy stored in the reactor in the transient state interval and to restore or discharge the magnetic energy to the DC source terminal of the GTO inverter in the steady state in both the motor-driving and -braking operation, and the third function to absorb or charge the motor kinetic energy and to regenerate or discharge the energy to the AC source side in the motor-braking operation.

Hereupon, the above surge voltage absorbing function is greatly dependent upon the voltage e_(c1) developed across the capacitor C₁. This voltage e_(c1) can be controlled by adjusting the off-time interval τ of the GTOs G₇ and G₈.

FIG. 8 shows the relationship between the off-time interval τ of the GTOs G₇ and G₈ and the capacitor voltage e_(c1) with the capacitance C₁ as parameter at a fixed dc source voltage Ed and a fixed frequency f of the GTO inverter 3. This graphical representation indicates that the longer the off-time interval τ of GTOs G₇ and G₈, the higher the voltage e_(c1), because the discharge time interval of the capacitor C₁ decreases. However, there exists little influence of the capacity of the capacitor C₁ upon the capacitor voltage e_(c1).

FIG. 9 shows the relationship between the frequency f of the GTO inverter 3 and the capacitor voltage e_(c1) with the off-time interval τ as parameter at a fixed dc source voltage Ed and a fixed capacitance C₁. This graphical representation indicates that the higher the frequency f, the higher the voltage e_(c1), because the commutation energy per unit time increases. However, the shorter the off-time interval τ, the capacitor voltage e_(c1) undergoes little influence of change in the frequency f.

FIG. 10 shows the relationship between the dc source voltage Ed and the capacitor voltage e_(c1) at a fixed frequency f of the inverter 3, a fixed off-time interval τ of the GTOs G₇ and G₈ and a fixed capacitance C₁. This graphical representation indicates that the capacitor voltage e_(c1) increases with increasing dc source voltage Ed and further e_(c1) always exceeds Ed.

As explained above, the off-time interval τ of the GTOs G₇ and G₈ incorporated in the surge voltage clamping circuit 6 according to the present invention exerts a serious influence upon the clamping operation of surge voltage.

Further, in the case where the DC source voltage Ed is low, it should be noted that since the capacitor voltage e_(c1) increases with increasing the off-time interval τ as depicted in FIG. 8, the clamping operation is deteriorated, thus resulting in sharp waveform change in the motor load current.

FIG. 11(A) shows an equivalent circuit diagram corresponding to the current path shown by thicker lines in FIG. 5(A). That is, FIG. 11(A) shows the steady state where both GTOs G₁ and G₆ are turned on. However, a discharging loop of the capacitor C₁ is neglected. FIG. 11(B) shows an equivalent circuit diagram coresponding to the current paths shown by thicker lines in FIG. 5(B). That is, FIG. 11(B) shows the transient state GTO G₁ has been turned off and GTO G₃ has been turned on with GTO G₆ kept on.

With reference to FIG. 11(A), it is possible to obtain a steady-state circuit equation as follows:

    (L.sub.d +2L)dI.sub.d /dt+(R.sub.d +2R.sub.1)i.sub.d +V.sub.u -V.sub.w =E.sub.d

    V.sub.u =E sin (ωt+ψ.sub.1 2π/3)

    V.sub.w =E sin (ωt+ψ.sub.1 -2π/3)

Similarly, with reference to FIG. 11(B), it is possible to obtain transient-state circuit equations as follows: ##EQU1## where R₁ : induction motor stator resistance

L: sum of stator and rotor leakage inductance

V_(u), V_(v), V_(w) : motor's counter electromotive force (CEMF) generated by fundamental component of input current

e_(c1) : initial capacitor voltage

E: peak phase voltage of motor CEMF

ψ: phase angle between fundamental component of input current and fundamental component of CEMF

V_(u) ', V_(v) ', V_(w) ': each phase terminal voltage

ω: inverter angular frequency.

I_(d) : dc current flowing from dc source Ed

i_(d1) : current flowing through phase u

i_(d2) : current flowing through phase v

i_(d3) : current flowing through diode D₆

E_(d) : average voltage of dc source Ed

R_(d) : resistance of dc reactor 7

L_(d) : inductance of dc reactor 7

FIG. 12 shows a timing chart of waveforms of the inverter shown in FIG. 4, in which the time interval enclosed between two dashed lines roughly corresponds to FIG. 2. In FIG. 12, the output current i_(u) rises and falls relatively gradually and surge voltage are sufficiently clamped or suppressed. Further, the output current i_(DBR) of the diode bridge-connected circuit 5 is generated for each commutation in pulsive waveform state to sequentially charge up the capacitor C₁.

The surge voltage clamping circuit according to the present invention has the following various features:

(1) The circuit operates stably when driving an induction motor at a high speed, because there are provided no vibration or commutation circuits to turn off ordinary thyristors.

(2) The circuit is high in energy conversion efficiency, because no commutating capacitors are provided.

(3) The circuit cost is reduced, because an ordinary DC reactor can be used in place of a cumulative DC reactors, because the energy stored in the capacitor C₁ is discharged frequently through the GTOs, and no commutating thyristors to generate sharp transient current are incorporated.

(4) The circuit can protect the GTOs (G₁ to G₆) of the inverter circuit from induced surge voltages in dependence upon relative-small GTOs (G₇ to G₁₀) of the clamping circuit.

(5) The magnitude of the surged voltage and the time interval of the transient state can be adjusted by adjusting the off-time intervals of the GTOs of the clamping circuit.

(6) The circuit can restore or regenerate surge voltage, magnetic or kinetic energy both in motor-driving operation and motor-braking operation.

It will be understood by those skilled in the art that the foregoing description is in terms of a preferred embodiment of the present invention wherein various changes and modifications may be made without departing from the spirit and scope of the invention, as set forth in the appended claims. 

What is claimed is:
 1. In a current type gate turn-off thyristor inverter having AC source terminals for driving an induction motor, comprising:(a) a GTO bridge-connected inverter; (b) a thyristor bridge-connected rectifier; (c) a diode bridge-connected commutation surge voltage rectifier connected to said GTO bridge-connected inverter; (d) a capacitor connected to said diode bridge-connected commutation surge voltage rectifier for storing a commutation surge voltage energy generated when each of the thyristors of said GTO bridge-connected inverter is turned off; (e) a cumulative reator having a first winding a positive side of which is connected to a positive terminal of said inverter and a second winding a negative side of which is connected to a negative terminal of said inverter; (f) a DC reactor connected between said rectifier and said inverter; (g) a first gate turn-off thyristor a cathode of which is connected to a negative side of the first winding of said cumulative reactor and an anode of which is connected to a positive side of said capacitor; (h) a second gate turn-off thyristor a cathode of which is connected to a negative side of said capacitor and an anode of which is connected to a positive side of the second winding of said cumulative reactor; (i) a first diode an anode of which is connected to the positive side of the second winding of said cumulative reactor and a cathode of which is connected to the positive side of said capacitor; and (j) a second diode an anode of which is connected to the negative side of said capacitor and a cathode of which is connected to the negative side of the first winding, the improvement wherein said first gate turn-off thyristor is turned on during steady state intervals of inverter commutation and off during transient state intervals of inverter commutation, and said second gate turn-off thyristor is turned on during steady state intervals of inverter commutation and off during transient state intervals of inverter commutation, and wherein said current type gate turn-off thyristor inverter further comprises: (k) a third gate turn-off thyristor an anode of which is connected to the positive side of said capacitor and a cathode of which is connected to the positive side of said second winding of said reactor, said third gate turn-off thyristor being turned on while the voltage charged in said capacitor exceeds a predetermined value; (l) a fourth gate turn-off thyristor an anode of which is connected to the negative side of said first winding of said reactor and a cathode of which is connected to the negative side of said capacitor, said fourth gate turn-off thyristor being turned on while the voltage charged in said capacitor exceeds a predetermined value; (m) a third diode an anode of which is connected to the negative side of said first winding of said reactor and a cathode of which is connected to the positive side of said capacitor; and (n) a fourth diode an anode of which is connected to the negative side of said capacitor and a cathode of which is connected to the positive side of said second winding of said reactor, whereby the commutation surge voltage energy stored in said capacitor during transient state of motor-driving operation is restored to the positive and negative terminals of said inverter through said first and second gate turn-off thyristors during steady state intervals of inverter commutation, magnetic energy stored in said reactor during transient state of motor-driving operation is recharged to said capacitor through said diode bridge-connected commutation surge voltage rectifier after said first and second gate turn-off thyristors have been turned off, the motor kinetic energy generated during motor-braking operation is stored in said capacitor through said first and second diodes and regenerated to the AC source terminal of said inverter through said third and fourth gate turn-off thyristors when the voltage across said capacitor exceeds a predetermined value, and magnetic energy stored in said reactor during motor-braking operation is recharged to said capacitor through said third and fourth diodes after the said third and fourth gate turn-off thyristors have been turned off. 